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Tuesday, 7 April 2009

Artificial cartilage

The smooth cartilage that covers the ends of long bones provides a level of lubrication that artificial alternatives haven't been able to rival – until now. Researchers say their lubricating layers of "molecular brushes" can outperform nature under the highest pressures encountered within joints, with potentially important implications for joint replacement surgery.

With every step we take, bones at the knee and hip rub against each other. That would quickly wear them away if it wasn't for the protection afforded by the thick layer of smooth and slippery cartilage that covers their ends.

No amount of polishing can remove all of the small imperfections from the stainless steel used in artificial joints. Any raised areas that are left grind against each other and release debris particles that soften the bone, explains Jacob Klein at the Weizmann Institute of Science in Rehovot, Israel.

Like bone, artificial joints must be covered with a cartilage-like layer. However, while it's possible to match cartilage's slick properties at lLinkow pressure, at the high pressures found in joints synthetic alternatives "seize up".

Battlefield lasers

"WE'RE doing our part to make gunpowder a 20th-century technology." So says Dan Wildt of Northrop Grumman, whose battlefield laser weapon passed another milestone last week.

In tests, it fired a 105-kilowatt beam - enough to destroy rockets, mortars and artillery shells - at a stationary target for 5 minutes. Unlike weapons such as Boeing's huge Airborne Laser, which burns chemical fuel, the solid-state laser consists of semiconductors that emit light when a voltage is applied. This makes them much smaller, allowing them to fit on the back of a "ruggedised" truck. They can also run on electricity from a diesel generator.

The laser weapon will ultimately pinpoint, track and destroy a rocket up to a couple of kilometres away, claims Northrop Grumman.

Vibrating touch screen puts Braille at the fingertips

OUCH-SCREEN devices like the iPhone are great when you can see them, but not much good if you are blind. Now a new way of presenting Braille characters on a mobile device could be the first step towards a Braille-ready touch-screen phone.

In Braille, letters are encoded using a two-by-three matrix in which each character is represented by a different configuration of raised and absent dots at the six locations. To display these dots on a touch-screen device, Jussi Rantala of the University of Tampere in Finland and colleagues used a Nokia 770 Internet Tablet, which has a piezoelectric material built into the touch screen that vibrates when an electric signal is applied to it. The team installed software that represents a raised dot as a single pulse of intense vibration, and an absent dot as a longer vibration made up of several weaker pulses (see diagram).

To discover how visually impaired volunteers would prefer to receive these vibrations, the team developed two different presentation methods. In the first, the user touches the screen on the left-hand side to read whether or not there is a bump in that position of the matrix, then moves their finger horizontally across the screen to read the remaining five dots. "But it wasn't that easy to read," says Rantala.

In the second method, the user simply places a finger anywhere on the screen and holds it still. The phone then displays a character by vibrating the sequence of six dots, each 360 milliseconds apart. "It took some time for them to start reading, because this representation is totally different from anything else that they had previously used," says Rantala. But once the volunteers were used to it, they were able to speed it up and read a character in as little as 1.25 seconds

The mobile future of the keyboard

Innovation is our new column that highlights the latest emerging technological ideas and where they may lead.

Cellphone users don't have it easy – many enter far more letters than numbers into their gadgets, but most phones still make you do so using a number pad. Meanwhile, the designers of smart phones seem determined to make touchscreen keyboards the norm before they have been fully perfected.

Although touch screens are growing in popularity with designers, tapping at images of buttons on a small, slippery surface does not provide a good user experience. Figuring out better ways to input text on touch screens is important for more than just phones too, as they become common in other places like desktop computers, gaming devices and coffee tables.

Recently some more innovative ideas have shown where the future of mobile touch-screen text input may lie. One that launched recently is Shapewriter – already available for the iPhone, Android and Windows Mobile devices.

It does away with the backwards-looking concept of pecking at images of keys on a glossy surface. A qwerty layout is still shown, but the user draws over it to link up the letters of a word they wish to write. The company behind Shapewriter says it has evidence this can be significantly faster than even a conventional touch keyboard – although at first glance, the shapes you draw even for relatively simple words seem elaborate.

Good vibrations

Another approach is to use a phone's vibrate function to give an uncannily real illusion of using physical buttons. Stephen Brewster's team at the University of Glasgow, UK, achieve the illusion with split-second pulses of vibrations chosen to provide sensations that feel like pressing a button, or shifting from key to key.

Next week at the Computer Human Interaction 2009 conference in Boston, Massachusetts, the team will present results of user tests on a Nokia N800 Internet Tablet equipped with the technology. The system uses the feedback to allow you to press harder on the screen for uppercase letters.

This video showing a mobile computer controlled using an interface projected onto any surface, like your hand, gives one possible view of the more distant future. But ultimately entering text may stop being a physical task altogether.

We reported last year on the first "voiceless phone call", placed using a neckband that makes it possible for someone to think words and have a computer read or type them outMovie Camera. The device detects and translates nerve signals sent by the brain to the vocal cords when we merely think about speaking a sound.

Made by Ambient Corporation it has the potential to combine the discreet silence of texting with the speed and accuracy of spoken language.

Their product is likely to help people with problems like motor neurone disease first, but they plan to develop it into a commercial product targeted as devices like phones too. However, before that happens we can expect to see many moLinkre attempts to perfect the mobile touch screen keyboard.

ntelligent car door clams up when danger's about

Accidents caused by car doors being opened into the path of oncoming vehicles or cyclists are common in cities. But these incidents could become a thing of the past, if doors that react to potential impacts catch on.

The brainchild of Michael Graf at BMW and Michael Strolz's team at the Technical University of Munich, the "haptic" – technology that gives tactile feedback – doors could cut both road injuries and repair bills, they say.

The current prototype looks like a normal car door, but an extra metal bar runs through its centre and connects to the car's frame between the hinges. In normal mode, the bar moves freely and doesn't affect the door's movement.

However, if sensors detect a nearby obstacle at the same time as an accelerometer detects an attempt to open the door, the door's swing is restricted by a linear motor attached to the bar.

To pass on more information to the user, the amount of door resistance is in proportion to the proximity of an object – for example, you might swing a door halfway open without problems before it gets stiffer as it nears a lamp post.

Mature technology

The current prototype uses ultrasonic sensors to spot dangers, but because they have a limited field of view, the next version will use cameras that can span 180°, says Strolz.

"Then we will be able to sense the complete workspace of the door and detect people walking by the car or cycling towards it," he says.

Reactions from 16 volunteers who tried the new door at BMW's Munich research centre have been encouraging, the team told a recent conference on haptics in Salt Lake City, Utah.

The technology is mature enough that a car factory could be pumping it out in cars within a year, says Strolz. The basic mechanism is like one already featured in some cars – for example, taxis with automatic passenger doors.Link

However, BMW is yet to make any decision on whether to roll it out.

ntelligent car door clams up when danger's about

Accidents caused by car doors being opened into the path of oncoming vehicles or cyclists are common in cities. But these incidents could become a thing of the past, if doors that react to potential impacts catch on.

The brainchild of Michael Graf at BMW and Michael Strolz's team at the Technical University of Munich, the "haptic" – technology that gives tactile feedback – doors could cut both road injuries and repair bills, they say.

The current prototype looks like a normal car door, but an extra metal bar runs through its centre and connects to the car's frame between the hinges. In normal mode, the bar moves freely and doesn't affect the door's movement.

However, if sensors detect a nearby obstacle at the same time as an accelerometer detects an attempt to open the door, the door's swing is restricted by a linear motor attached to the bar.

To pass on more information to the user, the amount of door resistance is in proportion to the proximity of an object – for example, you might swing a door halfway open without problems before it gets stiffer as it nears a lamp post.

Mature technology

The current prototype uses ultrasonic sensors to spot dangers, but because they have a limited field of view, the next version will use cameras that can span 180°, says Strolz.

"Then we will be able to sense the complete workspace of the door and detect people walking by the car or cycling towards it," he says.

Reactions from 16 volunteers who tried the new door at BMW's Munich research centre have been encouraging, the team told a recent conference on haptics in Salt Lake City, Utah.

The technology is mature enough that a car factory could be pumping it out in cars within a year, says Strolz. The basic mechanism is like one already featured in some cars – for example, taxis with automatic passenger doors.Link

However, BMW is yet to make any decision on whether to roll it out.

Google sees infrared in plan to scan world's books

THERE'S a hitch in Google's plan to digitise the world's books and make them searchable online: scanning them is taking too long.

That's because character recognition software needs a neat 2D image of the text. But book bindings cause pages to arch up either side of the spine - bending text and making it hard to interpret.

However, last week Google was granted a patent (US 7508978) on an answer to this problem. Its trick is to project an infrared pattern onto the open page spread. This lets a pair of infrared cameras map the three-dimensional shape of the pages by detecting distortion to the pattern. This in turn allows the distortion of the text to be determined - and therefore the degree of correction needed to read it accurately.

Could quantum mathematics shake up Google

MATHEMATICAL technique for studying disorder in quantum systems could improve internet keyword searches. It is able to spot significant patterns in large data sets such as web pages and text documents, and may even be adaptable to genome analysis.

Standard keyword searching compares word frequencies in one document with the frequencies in a standard corpus of text from many sources. If a word in the document occurs more frequently than average, it is considered important.

The new method gauges the importance of words in a document based on where they appear, rather than simply on how often they occur. "You should be able to detect an intrinsic property of a book without the need to compare it with different books," says Pedro Carpena, a physicist at the University of Malaga in Spain.

The importance of words in a document can be based on where they appear, rather than on how often

Carpena previously used mathematics from a field called random matrix theory to analyse quantum systems. He now says the same technique can be used to identify salient words in documents (Physical Review E, vol 79, p 035102).

Important words tend to be clustered together, Carpena says, while less important words appear more randomly distributed. This makes intuitive sense, he adds: as authors develop important ideas, they are likely to use relevant words many times in the same paragraph or page before moving on to other ideas. Less important words such as "and" and "but" tend to occur more evenly through the text.

The technique has worked well in Carpena's tests. Using random matrix theory to extract keywords from a book by Albert Einstein called Relativity: The special and general theory, he found "universe", "field", "gravitational", and "energy" among the top 10 results.

The method could even generate useful keywords when Carpena removed the spaces from a text document and asked the computer to identify significant letter combinations of between 2 and 35 characters long. This suggests it might also work on more abstract data sets. Carpena and his colleagues are currently testing the idea on the human genome to see whether it can extract useful information about genes.

It's not clear whether the search method is actually superior to existing ones, says Oren Etzioni, a computer scientist at the University of Washington in Seattle. He points out that Carpena has yet to compare his results with existing methods.

"Often great discoveries are made when techniques from one discipline are tried in another. This is potentially very promising, but they're wading into a crowded field," Etzioni says.

Could quantum mathematics shake up Google

MATHEMATICAL technique for studying disorder in quantum systems could improve internet keyword searches. It is able to spot significant patterns in large data sets such as web pages and text documents, and may even be adaptable to genome analysis.

Standard keyword searching compares word frequencies in one document with the frequencies in a standard corpus of text from many sources. If a word in the document occurs more frequently than average, it is considered important.

The new method gauges the importance of words in a document based on where they appear, rather than simply on how often they occur. "You should be able to detect an intrinsic property of a book without the need to compare it with different books," says Pedro Carpena, a physicist at the University of Malaga in Spain.

The importance of words in a document can be based on where they appear, rather than on how often

Carpena previously used mathematics from a field called random matrix theory to analyse quantum systems. He now says the same technique can be used to identify salient words in documents (Physical Review E, vol 79, p 035102).

Important words tend to be clustered together, Carpena says, while less important words appear more randomly distributed. This makes intuitive sense, he adds: as authors develop important ideas, they are likely to use relevant words many times in the same paragraph or page before moving on to other ideas. Less important words such as "and" and "but" tend to occur more evenly through the text.

The technique has worked well in Carpena's tests. Using random matrix theory to extract keywords from a book by Albert Einstein called Relativity: The special and general theory, he found "universe", "field", "gravitational", and "energy" among the top 10 results.

The method could even generate useful keywords when Carpena removed the spaces from a text document and asked the computer to identify significant letter combinations of between 2 and 35 characters long. This suggests it might also work on more abstract data sets. Carpena and his colleagues are currently testing the idea on the human genome to see whether it can extract useful information about genes.

It's not clear whether the search method is actually superior to existing ones, says Oren Etzioni, a computer scientist at the University of Washington in Seattle. He points out that Carpena has yet to compare his results with existing methods.

"Often great discoveries are made when techniques from one discipline are tried in another. This is potentially very promising, but they're wading into a crowded field," Etzioni says.

Russian rocket plans may prompt new space race

Russia is embarking on its most ambitious space project since the heady days of the space race: planning a new spaceship, launcher and even a new launch site. The plans are remarkably similar to NASA's Orion project and could provide a vital fallback if the US, Russia or other nations run into trouble with space missions.

Until now, Russia (and formerly, the Soviet Union) has simply upgraded its existing space facilities and hardware. Its three-person Soyuz spacecraft, for example, is now in its fifth generation in 40 years. But with top-level Kremlin backing, the Russian space agency Roskosmos is planning to entirely replace its current launch facilities, the rockets used to reach orbit, and the Soyuz itself. Future launches will take place from a new site near the Pacific coast city of Vladivostok.

"Post-Soviet Russia has never had a massive project of this kind," boasted Aleksey Krasnov, head of the agency's human spaceflight programme in a recent press briefing.

Today the agency is set to announce the contractor that will receive 800 million rubles ($24 million) to design the 20-tonne Soyuz reLinkplacement by June 2010. The first manned flight is planned for 2018.


Friday, 3 April 2009

scada

SCADA is an acronym that stands for Supervisory Control and Data Acquisition. SCADA refers to a system that collects data from various sensors at a factory, plant or in other remote locations and then sends this data to a central computer which then manages and controls the data

SCADA systems are used not only in industrial processes: e.g. steel making, power generation (conventional and nuclear) and distribution, chemistry, but also in some experimental facilities such as nuclear fusion. The size of such plants range from a few 1000 to several 10 thousands input/output (I/O) channels. However, SCADA systems evolve rapidly and are now penetrating the market of plants with a number of I/O channels of several 100 K: we know of two cases of near to 1 M I/O channels currently under development.

There are many parts of a working SCADA system. A SCADA system usually includes signal hardware (input and output), controllers, networks, user interface (HMI), communications equipment and software. All together, the term SCADA refers to the entire central system. The central system usually monitors data from various sensors that are either in close proximity or off site (sometimes miles away).

An industrial measurement and control system consisting of a central host or master (usually called a master station, master terminal unit or MTU); one or more field data gathering and control units or remotes (usually called remote stations, remote terminal units, or RTU's); and a collection of standard and/or custom software used to monitor and control remotely located field data elements. Contemporary SCADA systems exhibit predominantly open-loop control characteristics and utilise predominantly long distance communications, although some elements of closed-loop control and/or short distance communications may also be present.

Systems similar to SCADA systems are routinely seen in factories, treatment plants etc. These are often referred to as Distributed Control Systems (DCS). They have similar functions to SCADA systems, but the field data gathering or control units are usually located within a more confined area. Communications may be via a local area network (LAN), and will normally be reliable and high speed. A DCS system usually employs significant amounts of closed loop control.

SCADA systems on the other hand generally cover larger geographic areas, and rely on a variety of communications systems that are normally less reliable than a LAN. Closed loop control in this situation is less desirable.

Active Receiving Antennas

An active antenna is sometimes used for receiving purposes in instances where a normal antenna would be impossible to accommodate in a physical sense. Such an antenna is sometimes called an aperiodic antenna other people refer to it as an antenna booster.

Because modern receivers now cover such a broad frequency range it is also desirable to have a broad band antenna. In this tutorial I will attempt to address some of the theoretical justifications for using such a booster antenna.

It must be borne in mind at all times that an active receiving antenna is quite capable of introducing more severe problems into a receiving system than those which it is intended to solve.

Consider now some theoretical basics where we might compare a one metre long whip with a standard quarter wave antenna in the amateur 40 metre band. I simply selected the 40 metre band purely on whimsy and the principles could still apply at 80 metres, the a.m. radio band or even at long wave 175 Khz (1700 metres).

Conveniently a standard 1/4 wave antenna at 40 metres would measure approximately 10 metres in length. Assuming a distant signal, when received on that antenna, had a field strength of 10 uV per metre it would induce in our 1/4 wave antenna an open circuit voltage of 100 uV. Again for convenience only we will assume a pipe diameter of 20 mm.

Now on our one metre whip the same signal would induce an open circuit voltage of 10 uV. Are you following me here? I haven't used any fancy maths yet but I'll introduce some useful formulae now. NOTE the terms "an open circuit voltage", that is VERY significant. Here we will assume a whip diameter of 3 mm.

Whip Antenna Capacitance

whip antenna capacitance
Fig 1.

Where Ca is the antenna capacitance (in pF), 'h' is the height and 'a' is the whip diameter, both measured in metres. NOTE the co-efficient 0.7353 becomes 0.615 if dimensions are measured in inches. Ca is an approximation, many other factors come into play.

Short Whip Antenna Radiation Resistance

whip antenna radiation resistance
Fig 2.

Where Rr is the short whip antenna radiation resistance, 'h' is the height and l is the wavelength. This formula assumes a short vertical whip over a perfectly conducting plane which does not occur in reality.

Open Circuit Voltage

Open circuit voltage is simply the electric field strength multiplied by the physical height, as one example a particular signal might have a field strength of say 10 uV per metre (10 uV/M) and if the antenna height was say 12 metres long this would give us an open circuit voltage of 120 uV.

Simply it's the multiplication of 10 uV/M times the length. Be quite clear on the topic of open circuit voltage because it becomes quite important to your continued understanding.

Visualise a vertical antenna sitting out in the yard, 12 metres tall and conveniently for us there is only one signal available, it has a strength of 10 uV/M giving a total signal voltage on the antenna without any load attached of 120 uV. Close the eyes and think about it.

When a load is attached and we want to take some power from this antenna then that's when the fun starts. If you don't understand the concept of impedance then go back to my inelegant, but readily understandable analogy NOW because without it you're going to labour after this and probably miss the point.

Comparison of Antennas

The quarter wave antenna (with 100 uV open circuit voltage) is well known to have a radiation resistance of around 30 ohms (I've used 36 ohms in my sums below) and, disregarding matching considerations if terminated in a traditional 50 ohm load would form a voltage divider action as in fig. 3. This is purely for illustrative and comparative purposes only, please understand that particular point.

antenna voltage division - quarter wave antenna

Fig 3.

In practice of course we would use ground radials and utilize matching techniques. How is the reduction in open circuit voltage worked out? Well consider this:

Vout = Vin X [ RL / ( Rr + RL )] or;

58 uV = 100 uV X [ 50 / ( 36 + 50 )]

Let's look at our miserable one metre long whip which might be a few millimetres in diameter, we'll say for this exercise it is 3 mm diameter.

Using the formula above we get a Ca of about 12 pF (in practice it would most likely be more). At a nominal frequency of 7.5 Mhz the 12 pF reactance becomes a rough impedance of 1768 ohms in series with an insignificant Rr of much less than 1 ohm.

antenna voltage division  - active antenna

Fig 4.

Consider again,

Vout = Vin X [ RL / ( Z + RL )] or;

0.275 uV = 10 uV X [ 50 / ( 1768 + 50 )]

Essentially the 50 ohm load becomes almost a short circuit to the signal. Now I can assure you I have taken a considerable number of liberties here in the name of simplicity but the picture I have demonstrated is pretty much the real world situation.

In fact if you want to play around with receiving antennas I'd suggest you thoroughly digest this lesson as set out above because it should prove sobering (no pun intended).

Role of the Active Receiving Antenna

Now here is the general justification of an active antenna. If a high impedance load is connected to our one metre whip instead of the 50 ohm dead short then our calculations proceed as follows.

Assuming we use a field effect transistor as the amplifying device. The circuit configuration will be a source follower. A source follower exhibits high input impedance and relatively low output impedance. It also has a voltage LOSS.

This voltage loss (without going into a lot of theory) is about 10% BUT the power gain is almost infinite.

That statement will cause a great deal of confusion so consider this (purely hypothetical), if the input to our source follower is 10 uV into an impedance of 100,000 ohms (nominal for illustrative purposes) the input power taken by the FET is E2 / R which by way of calculation is 1 X 10 -15 watts.

The output delivered to a 200 ohm (nominal for illustrative purposes) load would be 90% of the input voltage (10% loss) or 9 uV and E2 / R which by way of calculation is 4.05 X 10 -13 watts or 405 times the input power or a power gain of 26 dB.

That is the role of the active device in this antenna. Simple!, well NO, there are quite a number of problems involved here. Firstly all FETS have some inherent input capacitance and this is one of the limiting problems. Assume a possible input capacitance of about 5 pF and a real world one meter whip having a capacitance of say 25 pF including holder, feed connection etc. Oh dear here comes that fink voltage divider action again.

The FET driving voltage is:

Vin / [ 1 + ( Ct / Ca )]

where Ct is the FET input capacitance (5 pF) and Ca is the antenna capacitance (all up 25 pF). If our one meter whip produces our former 10 uV open circuit voltage then the FET driving voltage is: 10 uV / [ 1 + ( 5 pF / 25 pF )] or 8.3 uV.

This reduction in voltage is nothing to become paranoid about because subsequent amplification will quite readily make up the difference. BTW the FET source follower configuration used here is sometimes considered to be an impedance converter.

Noise considerations

The first critical issue is one of S / N ratio. Assuming we are using a fairly good FET it should not degrade the overall noise performance of the receiving system. Below say 15 Mhz this is rarely an issue.

In the example of the quarter wave antenna and a 10 uV/M signal, external noise levels might be 1 uV/M.

These noise levels would include QRN (natural noise) and QRM (man made noise, such as my computer is presently doing to a nearby receiver). The noise is a constant ratio compared to received signal.

Whether we use a full quarter wave antenna or a physically short whip, the signal level and the noise level are going to be both proportional to physical length. The only consideration is whether the output signal level is going to be below the inherent receiver noise.

Limitations

So far you might imagine we could get away with an active receiving antenna as small as 50 mm in height (by the way it has been experimentally built). Unfortunately several vexing problems jump in our way. Don't discard the idea of an active receiving antenna because of them but be aware of the potential limitations. Some of these are:

1. Compromise of Receiver Dynamic Range:

Dynamic range might generally be described as the ratio of the level of strong out-of-band signals to the level of the weakest acceptable desired signal. An active receiving antenna is very broad band by nature and by virtue of its design. Indeed that often is the principle goal. Unfortunately it will also likely compromise the dynamic range of an otherwise excellent receiver because ALL signals present on the antenna are amplified equally.

I spoke earlier of signal levels of 10 uV/M, QRP-CW (low power morse code) enthusiasts wouldn't consider that much of a challenge. BUT with a broad band antenna don't be surprised to encounter signal levels of VOLTS per metre (not micro volts) from nearby transmitters especially strong stations in the A.M. radio band.

Many years ago I was fooling around with a basic crystal set tuned to a local A.M. station. I had a ferrite rod antenna which was tuned with a variable capacitor. NO external antenna was connected but I did have my high impedance oscilloscope attached to the ferrite rod. I was astounded to see a perfect A.M. signal of nearly 20V P/P.

2. Cross Modulation Distortion:

This occurs when the modulation (e.g. music) of an unwanted strong signal is transferred to a wanted weak signal (e.g. voice). I only highlighted those particular examples to give you an extreme understanding.

3. Inter modulation Distortion (IMD)

Receivers with many active devices, especially a receiver which has as the first active device an active receiving antenna, will frequently react in ways that do not always agree with theory. IMD is a complex problem.

In the case of an active receiving antenna you must also consider that the active device is capable of functioning as a mixer. I could give you pages of mathematics indicating trignomatic identities involving the production of a number of components with difference frequencies.

One definition (for our purposes) might be: "IMD - occurs in any non linear device (our FET) when driven by a complex signal having more than one frequency (our broad band antenna signals of - from 10 Khz up to 300 Mhz or more). The resultant signals (our output) become distorted".

As only one example, the possible effects of the A.M. radio band on an active antenna:

Signal (A) 900 Khz; Signal (B) 1500 Khz; - both at a field strength of 1000 uV/M (easily common) and finally our desired Signal (C) 3900 Khz - it's a lowly 10 uV/M signal.

A possible mixing action in our FET (they make good mixers) might be;

2 X Sig (A) + Sig (B) = Sig (C) 0r

( 2 X 1500) + 900 = 3900 Khz

considering the high levels of A and B and the low level of desired signal C what hope do you think you have recovering C signal? Hey and I've only mentioned two interferers and only ONE mathematical combination from around millions of available signals and a hell of a lot of possible mathematical combinations. Want to weep?

What to do?

Well don't slash you wrists or go and play on the freeway, consider why you really wanted an active antenna in the first place. Was it because someone told you they were the "ants pants"? Do you have space limitations?

If you live on acres somewhere then I would be hard pressed to imagine why you would even consider an active antenna. On the other hand if you live in an apartment and have decided the way to go for you is monitoring LF bands then an active antenna is all you may be able to consider.

If however you were a keen short wave listener then I'd certainly consider trying to unobtrusively lay as much small gauge insulated wire around the bottom of skirting boards, beneath carpets, throughout rooms as you could possibly lay. Consider all solutions, the cheaper the better.

An active receiving antenna is not necessarily restricted to a short whip. I've seen them used in conjunction with both tuned and untuned loops particularly on the V.L.F. AND L.F. bands.

Because they are tuned loops don't think they are necessarily selective. Anyone who has waded through my tutorials on L.C. Filters has seen that myth blown out of the water, or more correctly, put into proper perspective.

Go back to the basics. What was the original goal? If you can lay out as much wire as possible, the odds are any impedance mismatch could be accommodated with a relatively inexpensive simple tuner.

You might still incur some losses but they are often tolerable.

Radio Telescope

Antenna for a Radio Telescope

For starters if you wanted something infinitely simple then I would erect, as the article suggests, a half wave dipole antenna for your radio telescope. At 30 Mhz (10 metres) the theoretical length of a wavelength is 285 / 30 = 9.5 metres. The derivation of the 285 is the speed of light (300,000,000 metres per second) X 95% (velocity factor) divided by 1,000,000 (because we are using Mhz or millions of cycles per second).

It follows of course a half wave dipole antenna would be 9.5 / 2 = 4.75 metres or as I prefer 4,750 mm. See figure 1 below

half wave dipole antenna for a radio telescope

Fig 1. Half wave dipole antenna for a radio telescope

Ideally your radio telescope's half wave antenna, which consists of two quarter wave sections of 2375 mm each, should be installed as high as possible above ground. Much will depend upon the available real estate, your locality and 101 other factors beyond your control. The width between the posts or trees or whatever supporting your antenna is largely immaterial as long as you can accommodate the full 4,750 mm length plus insulators. Try and buy egg shell type insulators but failing that use thick plastic, Plexiglas or whatever is non metallic. Wood would be fine excepting it will absorb water which will destroy its insulating properties. The end pieces can be nylon rope passing through pulleys so the antenna can be lowered for maintenance.

For the antenna wire itself I like to use what we call in Australia 2.5 mm2 earth wire. Originally it had green insulation (now green/yellow) which was less obtrusive visually than bare copper wire, resists corrosion and its diameter usually has sufficient mechanical strength for spans of this size.

Don't get too paranoid about precise millimetre dimensions because this is for a radio telescope receiving application NOT transmitting where ideal 100% efficiency counts. Similarly don't get paranoid about impedance. Listen to me - please read that again.

Unfortunately, at 30 Mhz the physical size of your antenna places some considerable constraints upon you.

Personally I'd like to construct a yagi antenna, complete with corner reflector, pointing directly at the Milky Way. Better yet would be two or more such antennas stacked horizontally and vertically (impressive). At 30 Mhz the physical dimensions definitely wouldn't suit your average domestic back yard. You would have an antenna occupying something like 14.25 metres ( 47' ) in height, width and length!

Ideally I'd look at frequencies around 300 - 400 Mhz or more so we can have practical dimensions but at the expense of more complicated electronics. That is beyond the immediate scope of this article. If sufficient interest is generated (read considerable) I'll investigate it further, but read my addendum of 22nd April, 2000 at the bottom of the topic- heh! heh!.

Radio Telescope Receiver

In the original article the radio telescope coaxial cable fed a typical short wave receiver of apparent reasonable quality. I have several problems with that concept.

Firstly our one goal is to detect and record noise emitted by the Milky Way. At 30 Mhz we are on the threshold of the VHF spectrum where the inherent noise of the receiver itself becomes a dominant and limiting factor.

Secondly our coaxial cable introduces some losses into the system. Thirdly our receiver, if sufficiently sensitive and stable, does not need to be particularly flash. In short I wouldn't part with large sums of money on a suitable receiver when items 1 and 2 are limiting factors.

Remember these statements are applicable at VHF and above, possibly even down to 15 Mhz on occasion. Below 15 Mhz it's totally irrelevant.

Anticipating an inevitable question which will arise - "can I use an FM receiver". Answer: NO! because an FM receiver's selling point is the exact opposite to our goal, it kills noise and I won't involve myself in a lengthy discussion here as to why or how.

Radio Telescope Antenna Preamplifier / Mixer

The way I would consider going and indeed the substance of this radio telescope project is the antenna preamplifier / mixer approach. Never heard of it? Consider this.

In the receiver mentioned in the original article (typical sw receiver for its time) if it was tuned to 30 Mhz (actually 29.98 Mhz but that's insignificant here) the local oscillator would perhaps be at 30.455 Mhz and when mixed with a 30 Mhz received signal a resulting standard IF signal of 455 Khz is amplified many, many times and then detected for AM. If this is all double dutch to you then later on go to my main page and a whole new education awaits you.

The same principle applies to a cheap AM transistor radio operating in the AM radio band 531 Khz to 1611 Khz or whatever.

My theory is this:

At the threshold of VHF a low noise preamplifier is obligatory to establish as low a noise figure as is possible for a radio telescope to overcome the limitations of inherent receiver noise. Ideally this preamplifier is situated AT the antenna, this philosophy happens to be a little quirk of mine.

To overcome the losses of the coaxial cable etc. why not put the first mixer and other bits and pieces directly at the antenna also? Another little quirk!

We need the radio telescope preamplifier to establish a low noise figure and provide sufficient gain to overcome the insertion losses of the input filtering.

A double balanced diode mixer serves us well in these circumstances (see reference later)

A post mixer amplifier accomplishes a number of goals (again see later)

The only downside is near lightning strikes often play havoc with the type circuitry in such a vulnerable position. You can't have everything and remember this project is a lot like life, a matter of compromises. I have toyed with the idea of installing a NE-2 neon bulb at the input between signal and ground to bleed off static build up but I still remain in two minds.

So what have we decided? In place of the centre insulator in Fig 1 we will now instead have a shielded box (preferably diecast variety). In it we will construct, ugly style, in various compartments, a low noise preamplifier, various passive pre-selection circuitry, a passive mixer, a broad band post mixer amplifier and some output circuits. Our D.C. power supply will be fed through the same coaxial cable that the signals come down. Real Flash eh!

Here is a block diagram. The whole radio telescope circuitry may be called a "converter".

Radio Telescope converter block diagram

Fig 2 - Radio Telescope converter block diagram

Here we have a balun at the input because our half wave antenna presents a balanced 50 ohm (nominal) input whereas the input low pass filter and the succeeding circuitry is unbalanced. Similarly a coaxial cable presents an unbalanced load.

Radio Telescope Low pass filter

In this case we present our radio telescope antenna input after the balun to a 30 Mhz (nominal) two stage low pass filter. This is followed by a low noise dual gate fet amplifier. After that we have several stages of band pass filtering which feeds a double balanced passive diode mixer. The other ports of this mixer receive our 36 Mhz crystal oscillator frequency and the remaining port is the 6 Mhz band pass (36 - 30 = 6) filter and post IF amplifier. My choice of L.O. and I.F. frequencies was to an extent influenced by the ready availability of cheap microprocessor crystals at around 36 Mhz plus a desire to keep the I.F. at about 20% of the received input signal for image rejection purposes. Another quirk!

The input is a two stage low pass filter, why? Well stage one is a bog standard butterworth 3 pole filter with a Q of 1 (nominal 50 ohms both ports in / out) BUT modified to present a trap at the image frequency (30 + 36 = 66). No big deal however it doesn't hurt because in some areas that frequency falls within certain active TV channels. ABC-TV Channel 2 in Sydney is 63 - 70 Mhz and quite strong. The second stage is again a 3 pole butterworth filter with a Q of 10 and designed to transform the nominal 50 ohms to 2000 ohms input of the dual gate mosfet amplifier.

Here it is (notice I have included relevant reactance's and values in a table below) - a further quirk:

Radio Telescope input low pass filter

Fig 3 - Radio Telescope input low pass filter

The resistors in Fig 3 above are depicted solely to indicate input / output impedance's. These resistors are NOT actual resistors used in the circuit. Component values for capacitors are nearest standard 5% value or a suitable parallel combination.

Radio Telescope Low Pass Filter Component
C1
C2
C3
C4
C5
L1
L2
Value
100 pF
27 pf
100 pF
130 pf
27 pF
0.2 uH
1.18 uH
Reactance
50
39.67
50
40.5
200
39.7
222.5

Having used standard capacitors and a nominal 30 Mhz for our Radio Telescope input low pass filter we can run into a few problems. As one example take capacitor C1 with a required reactance of 50 ohms. At 30 Mhz the actual value calculates out to 106 pF or just marginally higher than a 5% tolerance 100 pF standard capacitor. On the other hand 100 pF EXACTLY at 50 ohms reactance is a frequency of 31.83 Mhz.

Generalized Rule with low pass filters - use the next lowest standard value capacitor DOWN from those actually calculated if necessary. With inductors use the next LOWEST number of whole turns e.g. a calculated 10.6 turns - use 10 turns NOT 11.

In practice with the circuit above in Fig 3, C3 and C4 would often be replaced by one single capacitor (it has the same effect). In this case there is a 5% tolerance standard close to (100 + 130 = 230 pF) the required value. A capacitor of 220 pF fits our generalized rule. Also be aware that although capacitors are 5% tolerance they can also vary considerably with temperature variations.

If you don't entirely understand this then perhaps you should see my tutorial on low pass filters, hopefully you will come away with some better understanding.

Radio Telescope Low noise amplifier

For our radio telescope this is going to be the first critical part. Purely on whim I'm going to use a dual gate mosfet. Another reason is I have on hand quite a number of these devices so if I can ever build the unit I have whatever I need. I think my choice of a 40673 should still be readily available - I will provide sources of all parts later on if I see sufficient interest is generated. Other devices may be substituted. I make no particular claim that this circuit or the devices selected are necessarily optimum.

According to my RCA data book the 40673 offers these performance features:

  • superior cross-modulation performance and greater dynamic range than bipolar or single-gate Fets (this may be somewhat negated by newer devices with which I have no experience)

  • wide dynamic range permits large-signal handling before overload

  • excellent thermal stability

Additionally the device features

  • high unneutralized RF power gain

  • low VHF noise figure - typically 3.5 dB at 200 Mhz
Radio Telescope low noise amplifier

Fig 4 - Radio Telescope low noise amplifier


Radio Telescope low noise amplifier components
C6
C7
C8a
C8b
C9
C10
C11
Value
100 pF
6P8
47 pF
47 pf
2P7
82 pF
15 pF

Without going into a lot of technical discussion about the 2 pole band pass filter ( C7 to C11 and L4 and L5 ) let me say for those who like to follow these things, the centre design frequency, Fo is 30 Mhz. The theoretical design bandwidth is about 1 Mhz and using the recommended toroids the unloaded Q of the inductors should be about 165. The filter equivalent impedance calculates out to about 2530 ohms. C7 is calculated to match back to the 2200 ohm source and C11 is calculated to match the 2530 filter to the 50 ohm input of the mixer. C9 the principal coupling capacitor which influences the bandwidth was calculated at 2.72 pf.

You will note two possible oddities in the table above. Firstly C7 and C9 are possibly funny values to some people, tell me which you would find the easier to discern without possible mistake, 6.8 pF or 6P8 and 2.7 pF or 2P7. Could not 6.8 pf to some tired old eyes be taken as 68 pF? The capital "P" merely indicates a decimal point placer as well as pico-farads. Similarly I have used 82R, 220R, 2K2, 33K and 100K for resistors

The second oddity to some people might be the two identical capacitors for C8. At my peril, I did not want to introduce trimmer capacitors and as the resonator capacitance Co at 30 Mhz with 0.27 uH inductors you need 106 pF. C8 needs to be Co - C7 - C9 or 106 - 6.8 - 2.7 = 96.5 so I elected to split it in two. You could equally use one 100 pf capacitor. C10 was arrived at similarly ( Co - C9 - C11 ). There is also stray capacitance to take into account.

If you prefer changes C8 could be 82 pF fixed and C10 could be 68 pF fixed, each with a 20 pF trimmer in parallel. This is fine if you have available a 30 Mhz signal source, know how to peak simple filters and have some means of detection. I'm punting on the fact that for most readers this is sufficiently difficult as it is with very limited test resources they have available.

Double balanced mixer

Again there is already a tutorial on this if you need to know. In this instance I'm going to use a literal "black box" approach by recommending we use a Mini-Circuits SBL-1 mixer. Knowing the insides of it won't necessarily help you and to build it from four hot carrier diodes plus two baluns would probably cost you more anyway. Here are the connections. Note this is the bottom view, pins facing toward you

double balanced mixer

Fig 5 Double balanced mixer

Now that is pretty straight forward isn't it? In actual practice we will mount the SBL-1 on its back on a piece of blank circuit board. Note pins 1 - 7 and 2 - 8 are standard 0.1" (2.54 mm) pitch and pins 1 and 2 are standard 0.3" apart. The device is the same size as an 8 pin I.C. Mini Dip.

Crystal Oscillator

Again we are going to use something pretty bog standard here. Surplus microprocessor crystals are readily available and quite cheaply too.

crystal oscillator

Fig 6 Crystal oscillator

All of these circuits are just plain text book designs used in pretty well every manufacturers' application notes e.g. RCA, Motorola, National, Amidon, Mini-Circuits and others. Of course I have used classic butterworth filter design techniques for the filters.

Post mixer amplifier and output circuits

Here is where the fun starts

post mixer amplifier

Fig 7 Post mixer amplifier and filtering

Looking from the left hand side of the schematic where the coaxial socket is, here is the output which goes via coaxial cable to your receiver.

Going down the schematic you will note the 10 uF/ 25v electrolytic capacitor, 220 uH choke, a LM-340T-12 voltage regulator and associated by-pass capacitors. At the receiver end we are going to have circuitry which will inject 15V DC up the coaxial cable and here we are tapping off the 15V DC, passing it through a choke etc. to clean it up and then regulating it a second time on board down to 12V DC. Here is where our power supply comes from!. It came UP the coaxial cable.

Back to the left of the schematic I have taken the IF output from pins 3 and 4 of the SBL-1 mixer which needs to see a 50 ohm termination (quite important!). I think from memory I learnt my next trick from Dr. Ulrich Rhode DJ2LR in one of his many excellent contributions to the now defunct Ham Radio magazine in the 1970's.

To preserve this 50 ohm termination we pass the output to a "diplexer". A diplexer simply means our output from the mixer (which contains myriad's of signals) is going to take two different paths, one high pass ( 2 X 220 pf and L6) and the other low pass (L7 and 68 pF).

This diplexer consists firstly of a high pass filter (L6 and two 220 pF capacitors). L6 and each capacitor exhibits 50 ohms reactance at the cut off frequency of about 14.5 Mhz. Some would question my choice of cut off frequency here because traditionally three times the IF frequency is used i.e. 3 X 6 Mhz = 18 Mhz. Personally I wouldn't slash my wrists over it either way. If you prefer to stick with tradition then make L6 50 ohms at 18 Mhz (0.44 uH or 10 turns on T-50-6) and change the two capacitors to 180 pF each.

An important note on the toroids! The AL factor for T-50-6 is 40 while the T-37-10 is 25. In some instances I have deliberately used a reduced number of turns than those calculated, this is to allow for some stray capacitance which does exist between windings. Also be aware some toroids, even from the same batch will offer varying AL factors. Nothing is perfect!

The other part of the diplexer is a simple L network comprising L7 and the 68 pF capacitor (normally includes a trimmer) which forms a low pass filter at about 6.5 Mhz and transforms the impedance from 50 ohms to 2K2. This filter is terminated by the 2K2 resistor at the gate of the fet although I suspect it doesn't greatly help our noise figure.

The high pass filter is terminated in an absorptive termination of two parallel 100 ohm resistors which of course equals 50 ohms. I elected once again not to include the trimmer because I doubt many readers will have suitable test equipment. If you do have test equipment replace the 68 pF fixed with a 20 pF trimmer in parallel with a 56 pF fixed capacitor. Ideally I would have left the 68 pF fixed and made L7 variable (another quirk of mine) but that's a lot more difficult to achieve in this day and age. Boy it's exciting to see a test signal grow on an oscilloscope when tuning an inductor.

The amplifier is a re-run of the earlier preamplifier we used and comes from RCA literature. The output transformer with the 2K2 resistor presents a load of 2K2 to the amplifier and the 40:6 turns ratio ( 40 / 6 = 6.667) which is pretty close to the square root of 2200 / 50 which is 6.63. The reactance of the transformer using a FT-50-77 ferrite (NOT iron powder) toroid is about 130,000 ohms at 6 Mhz. The inductance is in the region of 3.5 mH (milli-henries) and it's a plain vanilla RF transformer.

Review so far (for the technically minded)

  • We have a dipole antenna cut for close to 30 Mhz. Minor variations in dimension would not be significant.

  • This feeds our initial 50 ohm low Q (1) low pass filter (LPF) at slightly above 30 Mhz which also contains a trap at 66 Mhz

  • The signal then goes to a further higher Q (10) LPF which also transforms the 50 ohms up to a nominal 2K2 input to our preamplifier.

  • The input termination for these input low pass filters is our 50 ohm Antenna.

  • The output of the preamplifier also sees a load of 2K2 thanks to the loading resistor across L3 and passes to a two pole butterworth band pass filter of about 1 Mhz wide centred around 30 Mhz. (29.5 - 30.5 Mhz). To achieve a narrower bandwidth would require a higher design Q and once the design Q becomes an appreciable fraction of the unloaded Q (Qu) of the inductors, L3 to L5, then the filter insertion loss becomes substantial. Design Q is about 33 and the inductor Qu (from the Amidon data book) is about 165.

Tables indicate at a ratio of about 5 for a two resonator filter, the insertion loss is about 3 dB (if I wasn't lazy I'd calculate it but what the hell). In any event the output should represent a 50 ohm source to the double balanced mixer.

On reviewing this section I decided to become "un-lazy" and present the insertion loss calculations for you:


IL = - 10 log ( 1 - QL/QU)2

= - 10 log ( 1 - 33 / 165)2 = - 10 log ( 0.8) 2 = - 1.938 dB or 2 dB loss
  • I deliberately refrained from designing any narrow band circuitry in the output of the post mixer amplifier because I am unsure what bandwidth was required. A filter with a design bandwidth of 200 Khz could easily be inserted here.

  • The input filtering (both sides of the preamplifier) probably accounts for a total insertion loss of about 5-6 dB. The preamplifier gain is probably about 13 dB and the succeeding double balanced mixer is another loss of about 7 dB, so our gains and losses are about square after coming out of the mixer.

  • The diplexer probably accounts for a couple of dB loss and assuming a similar gain of 13 dB from the post amplifier I reckon overall we have a net gain from antenna to coaxial socket of about 10 dB. I'm guessing a noise figure of between 3 - 5 dB (as I said earlier NOT an optimum design)

  • For the non-technical, an approximate net 10 dB gain means, in signal voltage terms, that a one micro volt signal on the antenna becomes 3.162 micro volts at the coaxial socket. Not exactly heart stopping eh? However we should have established a significantly improved receiver noise figure and here that is the name of the game or object of the whole exercise.

  • Note: - for the true "NON-BELIEVERS" - the main gain ALWAYS comes from the final IF stages later in the receiver, not here!. Repeat that after me! Therefore don't be a "non-believer" and be the first to tempt fate and my wrath by sending me an email asking "How can I beef up the front end gain?". YOU DON'T.

  • The first amplifier is only designed to establish an acceptable "noise figure" and its net gain in conjunction with the post mixer amplifier gain are simply expected to recoup the net losses elsewhere in our converter system including some anticipated losses in the coaxial cable.

Reservations (mine)

1. As I said earlier this is a far from optimum design, however I am hazy at best as to what the actual goal is (technically). I have no experience in this type of reception and what is to be expected. Obviously we are attempting to receive "noise" and measure it.

All my technical life in receiving I've been doing the exact opposite, trying to avoid or minimize noise. In this case I guess we have to differentiate between "good noise" that emanates from the Milky Way and "bad noise" which comes from terrestrial signals (local radio signals and interference etc.) and the inherent noise in our receiver set up.

Certainly I would expect the set up I have suggested above to be a significant marked improvement over a coaxial cable of indefinite length, attached to a dipole at one end and to a short wave receiver at the other end.

2. Also the chosen frequency of 30 Mhz being so close in proximity to both the CB band and the Amateur Radio 10 metre band also causes me considerable unease.

On reflection I have a fairly shrewd hunch this frequency was selected solely because it was the upper limit of the original author's short wave receiver and hopefully beyond terrestrial signals. In fact I am quite confident about that statement.

3. The chosen frequency does not lend itself to easily physically constructing a good directional antenna with high gain, excellent front-to-back ratio and good side lobe rejection. That's high-tech-speak for "it'd be nice to point it at the Milky Way and pick up nothing else apart from our expected noise from that source and perhaps E.T.'s phone calls being returned!"

Hey, this exercise was brought to you by a guy well versed in the technology of DX-TV reception. Used to pull in usable TV signals in Sydney from as far away as New Zealand and Brisbane and that's no small technical feat, especially as I lived in a valley. For me this exercise is like trying to answer the age-old question "how long is a piece of string".

Answer: Twice the distance from one end to the centre

On balance with these reservations I think I could make a case for a good project in the UHF region say 300 - 400 Mhz (or higher) instead of the limitations 30 Mhz brings to a radio telescope.

On the positive side

If you have got this far then your knowledge should have been somewhat improved. If so, then my time spent was worthwhile.

Completion

At the receiver end we need some circuitry to strip off the 6 Mhz signals, pass them to your receiver and inject our 15V DC up the coaxial cable to power the converter. Here is my suggestion.

Receiver input and power supply

Fig 8 Receiver input and power supply

Presumably this part of the circuitry will be installed indoors and connected to the converter at the antenna by a suitable length of 50 ohm coaxial cable (not 75 ohm TV type).

You will note to the bottom left of Fig 8 a 15V A.C. plug pack (wall wart) socket depicted. To avoid the use of transformers and dangerous exposed 120V or 240V A.C. wiring I always recommend using plug packs or wall warts as they are called in the USA and elsewhere. Please NOTE it is a 15V (or slightly higher) A.C. NOT D.C. The current rating will be whatever minimum size you can obtain. The power consumed here is absolutely negligible.

Next follows a bridge rectifier. I specified 1N4004 diodes only because they are cheap in Australia. Any silicon rectifying diodes will do or if you prefer, buy a cheap small bridge rectifier instead of the individual diodes. The diode depicted above the 15V regulator is there to protect the regulator. The 2200 uF capacitor is not sacred in value either but ensure the voltage rating does not go below 25V nor its value below say 1000 uF. The 10 uF capacitor value is not sacred but the comments about voltage rating still apply. The 0.1 uF capacitor should be as close as possible to the output of the regulator. Another one on the input wouldn't hurt.

The fuse depicted is COMPULSORY. Voltage rating is irrelevant and current rating is about as small as you can get, I'd suggest 100 mA maximum. The red led and associated 1K5 resistor are optional extras with the LED so mounted as to indicate power is on.

The 0.1 uF capacitor shown in the signal line between the two signal sockets (IN / OUT) is also COMPULSORY and its exact placement as depicted is CRITICAL. It blocks D.C. power from the receiver but allows radio signals to pass through.


Solary Energy Projects

Microcontroller Based Temperature Monitoring


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Wednesday, 1 April 2009

pic based DTMFPhone Dialer

The company I work for recently upgraded their phone system. They got a new IP phone system that can supposedly do everything - the new phones have big LCD screens with fancy menus, lots of different modes, PC connection for remote control of the phone, e-mailing voice messages, and tons of other features. All that said, there was a critical piece missing from the picture - a way to dial a stored number while already in a call. This is a feature I use often - I'm using a calling card to call home and I need to first dial the 800 number and then send my calling card pin number. Well, believe it or not, the cool new phones with all the fancy software couldn't do this and I had to switch back to pencil and paper! "We will include this feature in release 8.0 or possibly 8.5" was the response from the phone company. Yeah right, like I'm going to wait for 8.5!

So, here is the obvious solution - a dialer box that connects to the phone and can be used to store and dial phone numbers, pins, credit card numbers and what not. The circuit uses a NTE1690 DTMF dialer chip and a PIC16F690 microcontroller.

Because this is an IP phone and I cannot just send the DTMF tones over the line, the easiest place to plug in the box is between the phone and the handset. The box has two RJ9 ports at the back - one gets connected to the headset's port on the phone, the other to the headset itself. The dial tones are sent over the microphone line. This way, it works both on IP-phones, and on standard (old fashioned, non-IP) ones.

Here is the thing, assembled and put in an old network transceiver box:


Sure looks ugly, :-) A friend of mine likened it to something from Dr. Evil's hideout. All the better - this is the way I like it! Ugly or not, the dialer works perfectly and can dial at blazing speed - so fast that it sounds more like R2-D2 than a phone.

The headset from the phone is connected to one of the RJ9 ports under the "AUI PORT" label; the other port is connected with an extra headset cable to the phone body. The pins sticking out on the other side from the port formerly known as "10BASE-T" can be connected to Microchip's PICkit 2 programmer for updating the firmware without taking the box apart.

The dialer is operated this way:
  • Storing a number. The device can store up to 12 numbers, identified by a bank ID. The ID is one of the buttons 0-9, SEQ, or RST. To store a number, press ENT, followed by the bank ID button. After pressing the bank ID, the green LED will light up, showing you that the device is expecting the number. Enter the number sequence (up to 15 digits); the SEQ button corresponds to the *-key on a phone; the RST number is #. You can use the DEL or TKO buttons to insert a pause. When you finish press ENT to store the number, or ABT to discard the changes and leave the old number stored in this bank unchanged. The green LED will turn off at this point. Example: To store the number 555-1234 in bank 0, press ENT-0-5-5-5-1-2-3-4-ENT.
  • Dialing a stored number. Just press one of the 12 bank ID keys (0-9, SEQ, or RST). If there is a number stored in that bank the device will dial the number, otherwise nothing will happen. Example: To dial the 555-1234 stored number (see "Storing number" above) simply press the 0 button.
  • Copying a stored number from one bank to another. Moving stored numbers around is a useful feature that I have not seen in any off-the-shelf dialer. To copy a number, first press ABT. The LED will light prompting you for source bank ID. Press the source bank button. The LED will remain on, waiting for a target bank ID. Press the target bank button. The number stored in the source bank will be copied to the target bank. You could cancel the operation at any time by pressing ABT. Example: To copy the 555-1234 number stored in bank 0 to bank 3, press ABT-0-3. After this, pressing the 3 button will dial 555-1234.
  • Manually dialing a number. Press the DEL button. The LED will turn on. Now you can use the keypad to dial any number. The SEQ button corresponds to the *-key on a phone; the RST number is #. To exit manual dial mode, press the DEL button again (pressing ENT, TKO, or ABT button will exit manual dial mode too). The LED will turn off at this point. Example: To manually dial 555-9999 press DEL-5-5-5-9-9-9-9-DEL.
  • Changing the dial speed. The maximum reliable dial speed depends on the system that you are dialing into and the noise on the line. To change the dial speed press TKO. The green LED will light up, indicating that the dialer is waiting for you to select dial speed. Press a number button between 0 and 7. The new dial speed will be set, 0 being the lowest speed, and 7 - the highest. The green LED will turn off after the speed is selected. Example: To switch to the highest dialing speed press TKO-7.

Here are some notes, in case you want to build such a dialer:
  • It's not very likely you'll find exactly the same keyboard that I have (this one I got for $0.33 apiece from a great electronics store in Santa Clara, CA; check also my list of SF Bay Area electronic shops). In case your keyboard have different layout, the change is easy to make - simply adjust the K_something constants defined in the beginning of HandheldDialer.asm. Define your layout by changing the hex digits to match the corresponding button on your keyboard (compare the source and the images of the keyboard to figure out the numbering of the buttons).
  • The schematics doesn't show the power supply. I'm using internal supply with a 9V battery and a 7805 voltage stabilizer. A sample design can be find here. Or maybe you will chose to construct something with external power supply?
  • Handsets may use different microphones, and can be wired in a different way. The schematics diagram shows two ways to connect to the handset cable - use whichever works for you. Special thanks to the friendly guys from the sci.electronics.design newsgroup for the help with the alternative design.

Schematics, source, and additional pictures

burdette switcher

The advantages and disadvantages I see (compared to "simple" designs) are:

  1. Parts are easily available.
  2. Layout and construction methods are not critical.
  3. Design can easily be tinkered with to scale up or down, or adjust to your requirements.
  4. You can get any voltage or any group of voltages you need. If you have the right transformer, you can get +5V, -5V, and -12V from one inverter.
  5. Parts are larger. My transformer is about 1.5 x 1.5 x 2 inches. Caps are bigger too.
  6. Parts count is higher, though maybe not compared to 3 separate "simple" designs.
  7. Parts cost may be higher unless you have a good transformer on hand. But again, maybe not compared to 3 separate "simple" designs.
  8. Regulation is not as good.

The core of the 12V inverter consists of Q1, T1, D2, and C3. When Q1 is on, D2 will be reverse biased and current will begin to build in T1. When Q1 turns off, that current will be forced to flow through C3 and D2, which will now be forward biased. Due to the high inductance of T1, the circuit operates in continuous mode, which means that the current through T1 never drops to zero, but ramps up and down about an average. I haven't actually measured it, but I suspect in my prototype the variation is less than 5% of the average.

Q2 is used to drive Q1 and invert the output of the '555. Note that since the base drive current of Q1 is 100 mA, R2 will dissipate significant power. The calculation is: 0.1 x 0.1 x 120 = 1.2 watts. Multiply by the duty cycle which may be as high as 60 or 65%, and you get about 0.78 watts. R1 ensures that Q1 turns off quickly, minimizing power dissipation and improving efficiency.

U1 is the oscillator which provides the PWM signal to Q1. R4, R5, and C5 create a 2000 Hz signal of approximately 60% duty cycle. As the output voltage increases, the feedback reduces the duty cycle and therefore the power supplied to the output. The frequency is also somewhat increased.

D3, Q3 and Q4 provide the output voltage sense and feedback to alter the PWM duty cycle of U1. As the output voltage increases, D3 conducts and turns on Q4. Q3 mirrors the collector current of Q4 to the control pin of U1 and protects that pin from being pulled below ground, possibly damaging the IC.

The -5V output is derived from the tap on T1, and is regulated by the autotransformer nature of T1. This means that if pin 3 is at -12V, pin 2 will be at -5V. In other words, the voltage ratio of the outputs is the same as the turns ratio of T1. Other voltages could be derived similarly by using other taps on the transformer. This is the way most PC power supplies are built - one transformer, many taps, only one of which is explicitly regulated.

It should be noted that if the -12V supply is unloaded, the -5V supply will not be well regulated. A certain minimum load must be placed on the -12V supply. I don't know what this is, but it should not be very high (10-20% of maximum load should be plenty).

Possible modifications:

  1. Use a FET in place of Q1. This could provide much higher output currents. My prototype runs out of steam at 0.7A because I don't drive Q1 hard enough to get more current through it. However, this provides one advantage. When first powering up, the current in T1 can build fairly high before C3 is fully charged. If Q1 were able to provide unlimited current, T1 might saturate, and the current would suddenly rise very high, possibly destroying Q1 or blowing the fuse. Since Q1 is current limited by virtue of its low gain at high current, we have somewhat of a soft startup.
  2. Increase the frequency. I measured the switching time on Q1, and it is around 400nS. This suggests that frequencies of 50-100 kHz are not unreasonable. An iron core transformer would have very high losses at this frequency, so you'd need ferrite. Be warned that a small ferrite would be much more likely to saturate on powerup. If you are using a FET, beware! You might also want to replace D1 and D2 with high speed diodes. You can use smaller output caps, but beware that that often means lower ripple current ratings and higher ESR!
  3. Change the transformer. The large core of a 12VA iron transformer is way overkill, even at 2000 Hz. Unfortunately, when you go to a smaller core, you get smaller wire and lower current ratings. What would be ideal is a smaller core with fewer turns of bigger wire. What that means is a lower voltage output rating with the same current rating. Good luck. If I really wanted to reduce the transformer size, I'd get a Miller high current torroidal inductor from Digi-key and boost the frequency to around 50 kHz. If I wanted more than one output voltage, I'd wind more turns and taps on the thing. That's not likely to be hard, since they don't have lots of turns on them.
  4. Reduce output ripple. My prototype has about 100 mV peak to peak ripple. About 30 mV of this is due to the current, capacitance, frequency combination, while about 70 mV is due to current and ESR (1.4 A x 0.05 ohms = 0.07 V). Note that the peak to peak current in C3 is around twice the output current. Since it's nearly a square wave, the RMS current is about half the peak to peak value. You could also set the output voltage higher and use a linear regulator to get really clean output.
  5. Improve the voltage detector. What's here is kind of crude. Though it works reasonably well, a reference diode and op-amp might improve it a little. Beware of making the gain too high though. I initially had a 1K resistor in R8, and that caused oscillation when Q1 turns off - when Q1 turns off, output voltage increases suddenly due to current suddenly flowing through the ESR of C3, which turns on Q4, which pulls down U1 pin 5, which turns Q1 back on, which causes the output voltage to decrease. Reducing the ESR of C3 should minimize this problem.
  6. Overload protection. If you overload this thing, you are likely to destroy Q1.

PIC Based Serial Port Servo Controller

PICServo Serial Port Servo Controller
PICServo Serial Port Servo Controller

Revision History

  • 28 Jan, 2002 - Bug found in 8 Servo motor variant, refer to 8 servo code page for details.
  • 28 Oct, 2001 - Serial Transfer Bug found.
  • Dec, 2005 - This project is no longer supported. The information is here for you to use. A new version of the PIC Servo controller is in the works and it will use a more modern PIC and support much faster serial communication and better position control commands.

Project Details

The PIC Based servo controller is a small project I designed to allow me to control some robotics. I then decided that I should publish it as it may be of interest to other people.

The Servo Controller uses a PIC16F84 microcontroller from Microchip to drive servo motors and digital outputs. It receives commands from a host computer via a standard RS232 serial interface.

You can download the PIC program which contains a HEX file which most PIC programmers can read; and a PIC Assembler code (ASM) file if you would like to see how it works or modify it.

I used the free PIC assembler from Microchip, the manufactures of the PIC range of Microcontrollers.

Several contributors have provided code samples that you can use to drive the PIC Servo project.

  • Nov, 2001 - Jose Maria Diaz de la Cruz of the Polytechnical University of Madrid (Spain) has provided an ActiveX control (ocx) and a Visual Basic Project. Note you will need to compile the Visual Basic project as it contains source only.
  • Feb 2002 - Richard Gardiner provided a Windows executable which is targeted at driving the 8 servo variant.

Here are some Acrobat files for the Schematic, Single sided PCB and Overlay artwork:

  • Schematic Diagram
  • PCB Viewed from Component Side
  • PCB Viewed from Copper Side
  • Component Overlay

You will need an Acrobat reader available for free from www.Adobe.com to view and print these files.

Note that the PCB is a single sided board, about 88 mm by 53 mm (3.5 inches by 2.1 inches). I have provided the track patters viewed from both sides of the board so you can choose the one that suits your method of production.

I have been asked for a version that will control eight servos, so I have altered the code to support this.

Servo Controller Commands

To control the servos and outputs we need to send commands to the PIC. Some of the commands are single byte commands; some however require two bytes.

The first byte always contains the command and the channel to which the command applies. We will call this the 'Command' byte. The second byte (when needed) will contain data for the command; we will call this the 'Data' byte and is used when we need to set the servo offset or position for example.

The command byte is split into two nibbles (4 bits), the upper one defines the command to execute and the lower defines the channel (which servo or digital output) is to be affected.

Bit 7Bit 6Bit 5Bit 4Bit 3Bit 2Bit 1Bit 0
CommandChannel Select

Currently, the software recognises only channels zero, one, two and three. The following are the commands and their numerical value. To build command bytes add the command value to the channel number.

HexDecimalData ByteDescription
000YesReset the controller. All outputs off, all servos disabled and all servos position and offset values are set to 128 (midrange). The data byte following this command MUST be zero. To ensure that a reset command it actually executed, send three consecutive zero bytes.
1016YesSet the servo output 'Position' value for the servo specified in the channel nibble. The data byte contains the new position value between 0 and 255.
2032YesSet the servo output 'Offset' value for the servo specified in the channel nibble. The data byte contains the new offset value between 0 and 255.
3048NoEnable Servo Output. Start generating the servo control pulse for the servo specified in the channel nibble.
4064NoDisable Servo Output. Stop generating the servo control pulse for the servo specified in the channel nibble.
5080NoSet Digital Output On (High). Set the digital output specified in the channel nibble high (5 volts).
6096NoSet Digital Output Off (Low). Set the digital output specified in the channel nibble low (0 volts).

Special consideration needs to be given to the reset command as the state of the command parser in the PIC is not known when it is issued we actually need to be a little tricky. The parser could be waiting for a command in which case it would work, or it could be waiting for a data byte in which case it would get a little confused.

Therefore we will make the reset command a series of three zeros and have the parser on the PIC detect that there is an extra byte and ignore it.

When the servo controller receives a reset command it will turn everything off, set all the servos position and offset values to midrange and send the string "Reset" to the serial port with a 'carriage return' and a 'new line' character appended.

Example Commands

Enable Servo 0: 48 + 0 = 48 (ASCII '0')
Enable Servo 2: 48 + 2 = 50 (ASCII '2')
Disable Servo 0: 64 + 0 = 64 (ASCII '@')
Disable Servo 2: 64 + 2 = 66 (ASCII 'B')
Set Servo 1 Position to 180: 16 + 1 = 17 then 180

Parts List

Resistors (Metal Film 1% or 5%, 0.25 W)
R11K
R2100K
Capacitors
C1, C2, C3, C4, C510uF 25 Volt Electrolytic
C647uF 25 Volt Electrolytic
C7, C822pF Ceramic
C9, C100.1uF MKT Polyester
Semiconductors
U1MAX232
U27805 voltage regulator
U3PIC16F84 programmed with SERVOCTL.HEX
D1IN4004
D25mm Red LED
Miscellaneous
Y14 MHz Crystal
83 pin PCB Header Pins. Buy a longer length and bread them up.
116 Pin IC Socket
118 Pin IC Socket
1DB9 90 degree female connector
12.5mm PCB Mount DC Power socket
3mm bolts, nuts and PCB standoffs for mounting

How R.C. Servo Motors Work

A servo motor consists of several main parts, the motor and gearbox, a position sensor, an error amplifier and motor driver and a circuit to decode the requested position. Figure 1 contains a block diagram of a typical servo motor unit.

The radio control receiver system (or other controller) generates a pulse of varying length approximately every 20 milliseconds. The pulse is normally between 1 and 2 milliseconds long. The length of the pulse is used by the servo to determine the position it should rotate to.

Servo Motor Block Diagram
Figure 1. Servo Motor Block Diagram

Starting from the control pulse we will work though each part of the diagram and explain how it all fits together. Once we have gone through how the servo works we will investigate how the control pulses can be generated with a microcontroller.

Pulse width to voltage converter

The control pulse is feed to a pulse width to voltage converter. This circuit charges a capacitor at a constant rate while the pulse is high. When the pulse goes low the charge on the capacitor is fed to the output via a suitable buffer amplifier. This essentially produces a voltage related to the length of the applied pulse.

The circuit is tuned to produce a useful voltage over a 1ms to 2ms period. The output voltage is buffered and so does not decay significantly between control pulses so the length of time between pulses is not critical.

Position Sensor

The current rotational position of the servo motor output shaft is read by a sensor. This is normally a potentiometer (variable resistor) which produces a voltage that is related to the absolute angle of the output shaft.

The position sensor then feeds its current value into the Error Amplifier which compares the current position with the commanded position from the pulse width to voltage converter.

Error Amplifier

The error amplifier is an operational amplifier with negative feedback. It will always try to minimise the difference between the inverting (negative) and non-inverting (positive) inputs by driving its output is the correct direction.

The output of the error amplifier is either a negative or positive voltage representing the difference between its inputs. The greater the difference the greater the voltage.

The error amplifier output is used to drive the motor; If it is positive the motor will turn in one direction, if negative the other. This allows the error amplifier to reduce the difference between its inputs (thus closing the negative feedback loop) and so make the servo go to the commanded position.

The servo normally contains a single integrated circuit and a hand full of discreet components to implement the entire control system.

Controlling a Servo Motor with a Microcontroller

From the above we can determine that we need to generate a pulse approximately every 20ms although the actual time between pulses is not critical. The pulse width however must be accurate to ensure that we can accurately set the position of the servo.

PWM modules

Many microcontrollers are equipped with PWM generators and most people initially consider using these to generate the control signals. Unfortunately they are not really suitable.

The problem is that we need a relatively accurate short pulse then a long delay; and generally you only have one PWM generator share between several servos which would require switching components outside the microcontroller and complicate the hardware.

The PWM generator is designed to generate an accurate pulse between 0% and 100% duty cycle, but we need something in the order of 5% to 10% duty cycle (1ms/20ms to 2ms/20ms). If a typical PWM generator is 8 or 10 bits say, then we can only use a small fraction of the bits to generate the pulse width we need and so we loose a lot of accuracy.

Timers

A more beneficial approach can be implemented with simple timers and software interrupts. The key is realising that we can run a timer at a faster rate and do a single servo at a time, followed by the next and the next etc. Each of the outputs is driven in turn for its required time and then turned off. Once all outputs have been driven, the cycle repeats.

This approach is demonstrated in the PIC servo controller project.

The timer is configured so that we have plenty of accuracy over the 1 to 2 millisecond pulse time. Each servo pin is driven high in turn and the timer configured to interrupt the processor when the pulse should be finished. The interrupt routine then drives the output low.

For simplicity, the output pins can be arranged on a single port and the value zero (0x00) written to the port to turn off all pins at once so that the interrupt routine does not need to know which servo output is currently active.

After the pulse has ended, the microprocessor sets up the next pulse and begins the process again.

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